Setting a transmission power level for a mobile unit

ABSTRACT

A spread signal is produced having an adjustable spread spectrum. A data signal is provided for transmission. The data signal is processed to have a first spread spectrum. The processed first spread spectrum data signal is filtered to have one out of a plurality of speed spectrums. The filtering capable of producing a signal having a spread spectrum of any of the plurality of spread spectrums. The filter one spread spectrum data signal is transmitted.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application No.12/712,657, filed Feb. 25, 2010, which issued on Jun. 14, 2011 as U.S.Pat. No. 7,961822, which is a continuation of U.S. patent applicationNo. 10/819,625, filed Apr. 7, 2004, which issued on Apr. 13, 2010 asU.S. Pat. No. 7,697,643, which is a continuation of U.S. patentapplication No. 10/071,971, filed Feb. 7, 2002, which issued on Apr. 13,2004 as U.S. Pat. No. 6, 721,350; which is a continuation of U.S. patentapplication No. 10/027,691, filed Oct. 19, 2001, which issued on Aug.26, 2003 as U.S. Pat. No. 6,611,548, which is a continuation of U.S.patent application No. 09/716,864, filed Nov. 20, 2000, which issued onDec. 11, 2001 as U.S. Pat. No. 6,330,272, which is a continuation ofU.S. patent application No. 09/277,400, filed Mar. 26, 1999, whichissued on Jan. 16, 2001 as U.S. Pat. No. 6,175,586, which is acontinuation of U.S. patent application No. 08/891,236, filed Jul. 10,1997, which issued on Nov. 30, 1999 as U.S. Pat. No. 5,995,538, which isa divisional of U.S. patent application no. 08/743,379, filed Nov. 4,1996, which issued on Nov. 10, 1998 as U.S. Pat. No. 5,835,527, which isa continuation of U.S. patent application No. 08/868,710, filed Jan. 4,1995, which issued on Nov. 12, 1996 as U.S. Pat. No. 5,574,747.

BACKGROUND

This invention relates to spread-spectrum communications, and moreparticularly to a multipath processor, variable bandwidth device, andpower control system.

Spread-spectrum modulation provides means for communicating in which aspread-spectrum signal occupies a bandwidth in excess of the minimumbandwidth necessary to send the same information. The band spread isaccomplished by modulating an information-data signal with achipping-sequence signal which is independent of an information-datasignal. The information-data signal may come from a data device such asa computer, or an analog device which outputs an analog signal which hasbeen digitized to an information-data signal, such as voice or video.The chipping-sequence signal is generated by a chip-code where the timeduration, T_(c), of each chip is substantially less than a data bit ordata symbol. A synchronized reception of the information-data signalwith the chipping-sequence signal at a receiver is used for despreadingthe spread-spectrum signal and subsequent recovery of data from thespread-spectrum signal.

Spread-spectrum modulation offers many advantages as a communicationssystem for an office or urban environment. These advantages includereducing intentional and unintentional interference, combating multipathproblems, and providing multiple access to a communications systemshared by multiple users. Commercially, these applications include, butare not limited to, local area networks for computers and personalcommunications networks for telephone, as well as other dataapplications.

A cellular communication network, using spread-spectrum modulation forcommunicating between a base station and a multiplicity of users,requires control of the power level of a particular mobile user station.Within a particular cell, a mobile station near the base station of thecell may be required to transmit with a power level less than thatrequired when the mobile station is near an outer perimeter of the cell.This adjustment in power level is done to ensure a constant power levelis received at the base station from each mobile station.

In a first geographical region, such as an urban environment, thecellular architecture may have small cells in which the respective basestation are close to each other, requiring a low power level from eachmobile user. In a second geographical region, such as a ruralenvironment, the cellular architecture may have large cells in which therespective base stations are spread apart, requiring a relatively highpower level from each mobile user. A mobile user who moves from thefirst geographical region to the second geographical region typicallyadjusts the power level of his transmitter in order to meet therequirements of a particular geographic region. If adjustments were notmade, a mobile user traveling from a sparsely populated region withlarger cells, using the relatively higher power level with hisspread-spectrum transmitter, to a densely populated region with manysmall cells may, without reducing the original power level of hisspread-spectrum transmitter, cause undesirable interference within thesmaller cell into which he has traveled and/or to adjacent cells. Also,if a mobile user moves behind a building and has his signal to the basestation blocked by the building, then the mobile user's power levelshould be increased. These adjustments must be made quickly, with highdynamic range and in a manner to ensure an almost constant receivedpower level with low root mean square error and peak deviations from theconstant level.

SUMMARY

A spread signal is produced having an adjustable spread spectrum. A datesignal is provided for transmission. The data signal is processed tohave a first spread spectrum. The processed first spread spectrum datasignal is filtered to have one out of a plurality of spread spectrums.The filtering capable of producing a signal having a spread spectrum ofany of the plurality of spread spectrums. the filtered one spreadspectrum data signal is transmitted.

BRIEF DESCRIPTION OF THE DRAWING(S)

The accompanying drawings, which are incorporated in and constitute apart of the specification, illustrate preferred embodiments of theinvention, and together with the description serve to explain theprinciples of the invention.

FIG. 1 illustrates channel impulse response giving rise to severalmultipath signals;

FIG. 2 illustrates conditions leading to two groups of several multipathsignals;

FIG. 3 is a block diagram of a multipath processor using two sets ofcorrelators for despreading a spread-spectrum signal received as twogroups of spread-spectrum signals;

FIG. 4 is a block diagram for generating chirping-sequence signals withdelays;

FIG. 5 is a tapped-delay line model of a communications channel;

FIG. 6 is a block diagram of a correlator;

FIG. 7 is an auto correlation function diagram generated from thecorrelator of FIG. 6;

FIG. 8 is a black diagram for tracking a received signal;

FIG. 9 is a block diagram for combining a pilot signal from a receivedspread-spectrum signal;

FIG. 10 is a block diagram for tracking a pilot signal embedded in apilot channel of a spread-spectrum signal;

FIG. 11 illustrates cross-correlation between a received signal and areferenced chipping-sequence signal, as a function of referenced delay;

FIG. 12 illustrates the center of gravity of the cross-correlationfunction of FIG. 11;

FIG. 13 is a block diagram of a multipath processor using two sets ofmatched filters for despreading a spread-spectrum signal received as twogroups of spread-spectrum signals;

FIG. 14 is a block diagram of a multipath processor using three sets ofcorrelators for despreading a spread-spectrum signal received as threegroups of spread-spectrum signals;

FIG. 15 is a block diagram of a multipath processor using three sets ofmatched filters for despreading a spread-spectrum signal received asthree groups of spread-spectrum signals:

FIG. 16 is a block diagram of a variable-bandwidth spread-spectrumdevice;

FIG. 17 illustrates chips of a spread-data signal;

FIG. 18 illustrates impulse signals corresponding to the chips of thespread-data signal of FIG. 17;

FIG. 19 is an alternative block diagram of the variable-bandwidthspread-spectrum device of FIG. 16;

FIG. 20 is a block diagram of a base station;

FIG. 21 is a block diagram of a mobile station;

FIG. 22 illustrates nonlinear power adjustment;

FIG. 23 illustrates linear and nonlinear cower adjustment;

FIG. 24 illustrates fades during transmission for multiple signals orequivalent power received at a base station;

FIG. 25 illustrates an adaptive power control signal of broadcast powerfor a fixed step algorithm;

FIG. 26 illustrates despread output power for a fixed step algorithm;

FIG. 27 illustrates an adaptive power control signal of broadcast powerfor a variable step algorithm; and

FIG. 28 illustrates despread output power for a variable step algorithm.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT(S)

Reference now is made in detail to the present preferred embodiments ofthe invention, examples of which are illustrated in the accompanyingdrawings, wherein like reference numerals indicate like elementsthroughout the several views.

In a multipath environment, a signal reflects from several buildings orother structures. The multiple reflections from the several buildingscan result in several signals, or several groups of signals, arriving ata receiver. FIG. 1 illustrates a signal arriving in time as severalsignals. FIG. 2 illustrates a signal arriving in time as two groups ofseveral signals. The multiple signals arriving at the receiver usuallydo not arrive with a uniform spread over time. Thus, in a multipathenvironment, a received signal r(t) may include two or more groups ofspread-spectrum signals.

In the multipath environment, a spread-spectrum signal is assumed togenerate a plurality of groups of spread-spectrum signals, with eachgroup having a plurality of spread-spectrum signals. The plurality ofgroups is the result of the spread-spectrum signal reflecting in amultipath environment. As a means of responding to and dealing with thisplurality of groups, the multipath processor is an improvement to aspread-spectrum receiver system. In the exemplary arrangement shown inFIG. 3, a multipath processor for tracking a spread-spectrum signal isshown. The multipath processor is used as part of a spread-spectrumreceiver system.

The multipath processor includes first despreading means, seconddespreading means, first combining means, second combining means, andselecting means or output-combining means. The first combining means iscoupled between the first despreading means and the selecting means orthe output-combining signal. The second combining means is coupledbetween the second despreading means and the selecting means or theoutput-combining means.

The first despreading means despreads a received signal having a firstplurality of spread-spectrum signals within a first group. The firstdespreading means thus generates a first plurality of despread signals.The first combining means combines, or adds together, the firstplurality of despread signals to generate a first combined-despreadsignal.

The second despreading means despreads the received signal having asecond plurality of spread-spectrum, signals within, a second group. Thesecond despreading means thereby generates a second plurality ofdespread signals. The second combining means combines, or adds together,the second plurality of despread signals as a second combined-despreadsignal.

The selecting means selects either the first combined-despread signal orthe second combined-despread signal. The selected combined-despreadsignal is outputted from the selecting means as an output-despreadsignal. The selecting means may operate responsive to the strongersignal strength of the first combined-despread signal and the secondcombined-despread signal, least mean square error, a maximum likelihood,or other selection criteria. Alternatively, using output-combining meansin place of selecting means, the outputs of the first combining meansand the second combining means may be coherently combined or addedtogether, after suitable weighting.

As shown in FIG. 3, the first despreading means may include a firstplurality of correlators for despreading, respectively, the firstplurality of spread-spectrum signals. The first plurality of correlatorsis illustrated, by way of example, as first multiplier 111, secondmultiplier 112, third multiplier 113, first filter 121, second filter122, third filter 123, first chipping-sequence signal g(t), secondchipping-sequence signal g(t-T_(o)) and third chipping-sequence signalg(t-2T_(o)). The second chipping-sequence signal g(t-T_(o)) and thethird chipping-sequence signal g(t-2T_(o)) are the same as the firstchipping-sequence signal g(t), but delayed by time T_(o) and time2T_(o), respectively. The delay between each chipping-sequence signal,preferably, is a fixed delay T_(o). At the input is received signalr(t). The first multiplier 111 is coupled between the input and thefirst filter 121, and to a source of the first chipping-sequence signalg(t). The second multiplier 112 is coupled between the input and thesecond filter 122, and to a source of the second chipping-sequencesignal g(t-T_(o)). The third multiplier 113 is coupled between the inputand the third filter 123, and to a source of the third chipping-sequencesignal g(t-2T_(o)). The outputs of the first filter 121, the secondfilter 122 and the third filter 123 are coupled to the first adder 120.

Circuitry and apparatus are well known in the art for generatingchipping-sequence signals with various delays. Referring to FIG. 4, achipping-sequence generator 401 is coupled to a voltage-controlledoscillator 402, and a plurality of delay devices 403, 404, 405,406. Thevoltage-controlled oscillator receives a group-delay signal. Thegroup-delay signal corresponds to the time delay that the group ofchipping-sequence signals used for despreading a particular group ofreceived signals. The voltage-controlled oscillator 402 generates anoscillator signal. The chipping-sequence generator 401 generates thefirst chipping-sequence signal g(t) from the oscillator signal, with aninitial position of the first chipping-sequence signal g(t) determinedfrom the group-delay signal. The first chipping-sequence signal g(t) isdelayed by the plurality of delay devices 403, 404, 405, 406, togenerate the second chipping-sequence signal g(t-T_(o)), the thirdchipping-sequence signal g(t-2T_(o)), the fourth chipping-sequencesignal (gt-3T_(o)), etc. Thus, the second chipping-sequence signalg(t-2T_(o)) and the third chipping-sequence signal g(t-2T_(o)) may begenerated as delayed versions of the first chipping-sequence signalg(t). Additionally, acquisition and tracking circuitry are part of thereceiver circuit for acquiring a particular chipping-sequence signalembedded in a received spread-spectrum signal.

Optionally, the multipath processor of FIG. 3 may include firstweighting device 131, second weighting device 132 and third weightingdevice 133. The first weighting device 131 is coupled to the output ofthe first filter 121, and a source of a first weighting signal W₁. Thesecond weighting device 132 is coupled to the output of the secondfilter 122, and to a source of the second weighting signal W₂. The thirdweighting device 133 is coupled to the output of the third filter 123and to a source of the third weighting signal W₃. The first weightingsignal W₁, the second weighting signal W₂ and the third weighting signalW₃ are optional, and may be preset within the first weighting device131, the second weighting device 132 and the third weighting device 133,respectively. Alternatively, the first weighting signal W₁, the secondweighting signal W₂, and the third weighting signal W₃ may be controlledby a processor or other control circuitry. The outputs of the firstfilter 121, the second filter 122, and the third filter 123 are coupledthrough the first weighting device 131, the second weighting device 132and the third weighting device 133, respectively, to the first adder120.

Similarly, the second despreading means may include a second pluralityof correlators for despreading the second plurality of spread-spectrumsignals. The second plurality of correlators is illustrated, by way ofexample, as fourth multiplier 114, fifth multiplier 115, sixthmultiplier 116, fourth filter 124, fifth filter 125, sixth filter 126,fourth chipping-sequence signal g(t-T_(D1)), fifth chipping-sequencesignal g(t-T_(o)—T_(D1)), and sixth chipping-sequence signalg(t-2T_(o)-T_(D1)). The fourth multiplier 114 is coupled between theinput, and the fourth filter 124, and a source of the fourthchipping-sequence signal g(t-T_(D1)). The fifth multiplier 115 iscoupled between the input and the fifth filter 125 and a source of thefifth chipping-sequence signal g(t-T_(o)-T_(D1)). The sixth multiplier116 is coupled between the input and the sixth filter 126, and a sourceof the sixth chipping-sequence signal g(t-2T_(o)-T_(D1)). The fourthchipping-sequence signal g(t-T_(D1)), the fifth chipping-sequence signalg(t-T_(o)-T_(D1)) and the sixth chipping-sequence signalg(t-2T_(o)-T_(D1)) are the same as the first chipping-sequence signalg(t), but delayed by time T_(D1) time T_(o)+T_(D1), and time2T_(o)+T_(D1), respectively. The second plurality of correlators therebygenerates the second plurality of despread signals. The outputs of thefourth filter 124, the fifth filter 125 and the sixth filter 126 arecoupled to the second adder 130.

At the output of the fourth filter 124, the fifth filter 125, and thesixth filter 126, optionally, may be fourth weighting device 134, fifthweighting device 135, and sixth weighting device 136. The fourthweighting device 134, fifth weighting device 135, and sixth weightingdevice 136 are coupled to a source which generates fourth weightingsignal W₄, fifth weighting signal W₅, and sixth weighting signal W₆,respectively. The fourth weighting signal W₄, the fifth weighting signalW₅, and the sixth weighting signal W₆ are optional, and may be presetwithin the fourth weighting device 134, the fifth weighting device 135.and the sixth weighting device 136, respectively. Alternatively, thefourth weighting signal W₄, the fifth weighting signal W₅, and the sixthweighting signal W₆ may be controlled by a processor or other controlcircuitry. The outputs of the fourth filter 124, fifth filter 125, andsixth filter 126 are coupled through the fourth weighting device 134,fifth weighting device, 135 and sixth weighting device 136,respectively, to the second adder 130. The output of the first adder 120and the second adder 130 are coupled to the decision device 150. Thedecision device 150 may be a selector or a combiner.

The weighting devices may be embodied as an amplifier or attenuationcircuits, which change the magnitude and phase. The amplifier orattenuation circuits may be implemented with analog devices or withdigital circuitry. The amplifier circuit or attenuation circuit may beadjustable, with the gain of the amplifier circuit or attenuationcircuit controlled by the weighting signal. The use of a weightingsignal with a particular weighting device is optional. A particularweighting device may be designed with a fixed weight or a preset amount,such as a fixed amount of amplifier gain.

FIG. 5 is a tapped-delay-line model of a communications channel. Asignal s(t) entering the communications channel passes through aplurality of delays 411, 412, 413,414, modeled with time T_(o). Thesignal s(t), for each delay, is attenuated 416, 417, 418 by a pluralityof complex attenuation factors h_(n−1), h_(n), h_(n+)and adder 419. TheOUTPUT from the adder 419 is the output from the communications channel.

A given communications channel has a frequency response which is theFourier transform of the impulse response,

$\begin{matrix}{\mspace{79mu} {{Equation}\mspace{14mu} 1}} & \; \\{\mspace{79mu} {{{H(f)} = {\text{?}a}},{{\text{?}.\text{?}}\text{indicates text missing or illegible when filed}}}} & i\end{matrix}$

where a_(i)represents the complex gains of the multipaths of thecommunications channel, and t_(i) represents the delays of themultipaths of the communications channel.

Consider the communications-channel-frequency response, H_(n)(f). Thecommunications-channel-frequency response has a band of interest, B.Hereafter, this band, of interest is fixed, and thecommunications-channel-frequency response H_(o)(f) is the equivalentlowpass filter function. The communications-channel-frequency responseexpands in Fourier series as

Equation 2

H

(ƒ)=Σ h _(n) e

  i.

where h_(n) represents Fourier coefficients. This is a tapped-delay-linemodel of the communications channel for which the receiver in FIG. 3acts as a matched filter when T_(o)=1/B, and the weights W_(n) are setto the complex conjugate of the values h_(n). That is, W_(n)=h_(n).

Preferably, each correlator of the first plurality of correlatorsdespreads with a chipping-sequence signal g(t) which has a time delaydifferent from each time delay of each chipping-sequence signal used,respectively, with each of the other correlators of the first pluralityof correlators. The first plurality of correlators useschipping-sequence signals g(t), g(t-T_(o)), g(t-2T_(o)), where T_(o) isthe time delay between chipping-sequence signals. The time delay T_(o)may be the same or different between each chipping-sequence signal. Forillustrative purposes, time delay T_(o) is assumed to be the same.

Similarly, each correlator of the second plurality of correlatorsdespreads with a chipping-sequence signal having a time delay differentfrom each time delay of each other chipping-sequence signal used,respectively, with each of the other correlators of the second pluralityof correlators. Also, each correlator of the second Plurality ofcorrelators despreads with a chipping-sequence signal having the timedelay T_(D1) different from each time delay of each chipping-sequencesignal used with each respective correlator of the first plurality ofcorrelators. Thus, the second plurality of correlators useschipping-sequence signals g(t-T_(D1)), g(t-T_(o)-T_(D1)),g(t-2T_(o)-T_(D1)) where time delay T_(D1) is the time delay between thefirst plurality of correlators and the second plurality of correlators.The time delay T_(D1) is also approximately the same time delay asbetween the first received group of spread-spectrum signal sand thesecond received group of spread-spectrum signals.

FIG. 6 illustrates a correlator, where an input signal s(t) ismultiplied by multiplier 674 by a delayed version of the input signals(t-T). The product of the two signals is filtered by the filter 675,and the output is the autocorrelation function R(T). The autocorrelationfunction R(T) for a square wave input signal s(t) is shown in FIG. 7.Over a chip time T_(c), the correlation function R(T) is maximized whenpoints A and B are equal in amplitude. A circuit which is well known inthe art for performing this function is shown in FIG. 8. In FIG. 8, thedespread signal s(t) is delayed by a half chip time T_(o/2), andforwarded by half a chip time T_(o/2). Each of the three signals ismultiplied by the received signal r(t). The outputs of the delayed andforwarded multiplied signals are filtered, and then amplitude detected.The two filtered signals are combined by subtracting the delayed versionfrom the forwarded version, and the difference or error signal is usedto adjust the timing of the chipping-sequence signal used to despreadsignal s(t). Accordingly, if the delayed version were ahead of theforwarded version, the chipping-sequence signal for despread signal s(t)would be delayed. Likewise, if the forwarded version were ahead of thedelayed version, then the chipping-sequence signal for despreadingsignal s(t) would be advanced. These techniques are well known in theart.

A similar technique is used for estimating a pilot signal from areceived signal r(t), which has passed through a multipath environment.Referring to FIG. 9, the lower part of the diagram shows correlatorscorresponding to the correlators previously shown in FIG. 3. The upperpart of the diagram shows the received signal processed by delayedversions or the pilot chipping-sequence signal g_(p)(t). In FIG. 9, thereceived signal r(t) is multiplied by the pilot signal g_(p)(t) and aplurality of delayed versions of the pilot signal g_(p) (t-T_(o)), . . ., g_(p) (t-kT_(o)) by a plurality of multipliers 661, 651,641. Theoutput of the plurality of multipliers 601, 651, 841, are each filteredby a plurality of filters 662, 652, 642, respectively. The output of theplurality of filters 662, 652, 642 are multiplied by a second pluralityof multipliers 663, 653, 643 and respectively filtered by a secondplurality of filters 664, 654, 644. The outputs of the second pluralityof filters 664, 654, 644 are processed through a plurality of complexconjugate devices 665, 655, 645. The outputs of the plurality of complexconjugate devices 665, 655, 645 are the plurality of weights W₁, W₂,W_(k), respectively. The plurality of weights are multiplied by theoutput of the first plurality of filters 662, 652, 642, by a thirdplurality of multipliers 666, 656, 646, and then combined by thecombiner 667. At the output of the combiner 667 is acombined-despread-pilot signal.

Each of the second plurality of pilot filters 664, 654, 644 has abandwidth which is approximately equal to the fading bandwidth. Thisbandwidth typically is very narrow, and may be on the order of severalhundred Hertz.

Referring to FIG. 10, the output of the combiner 667 is multiplied by afourth multiplier 668, and passed through an imaginary device 669 fordetermining the imaginary component of the complex signal from thefourth multiplier 668. The output of the imaginary device 669 passesthrough a loop filter 672 to a voltage controlled oscillator 673 or anumerically controlled oscillator (NCO). The output of the voltagecontrolled oscillator 673 passes to the fourth multiplier 668 and toeach or the second plurality of multipliers 663, 653, 643.

Referring to FIG. 11, the foregoing circuits can generate across-correlation function between the received signal and a referencedpilot-chipping signal as a function of referenced delay, or lag. Asshown in FIG. 11, these points of cross-correlation can have a center ofgravity. The center of gravity is determined when the left mass equalsthe right mass of the correlation function, as shown in FIG. 12. Acircuit, similar to that shown in FIG. 8, coupled to the output of thefourth multiplier 668, can be used for aligning a chipping-sequencesignal of the pilot channel.

As an alternative embodiment, as shown in FIG. 13, the first despreadingmeans may include a first plurality of matched filters for despreadingthe received signal r(t) having the first plurality of spread-spectrumsignals. At the output of the first plurality of matched filters is thefirst plurality oC despread signals. Each matched filter of the firstplurality of matched filters has an impulse response h(t), h(t-T_(o)),h(t-2T_(o)), etc., with a time delay T_(o) offset from the other matchedfilters. Referring to FIG. 13, by way of example, a first matched filter141 is coupled between the input and through the first weighting device131 to the first adder 120. A second matched filter 142 is coupledbetween the input and through the second weighting device 132 to thefirst adder 120. A third matched filter 143 is coupled between the inputand through the third weighting device 133 to the first adder 120. Asmentioned previously, the first weighting device 131, the secondweighting device 132, and the third weighting device 133 are optional.The first weighting device 131, the second weighting device 132, and thethird weighting device 133 generally are connected to a source of thefirst weighting signal W₁, the second weighting signal W₂, and the thirdweighting signal W₃, respectively. The first plurality of matchedfilters generates the first plurality of despread signals.

Similarly, the second bespreading means may include a second pluralityof matched filters for despreading the received signal r(t) having thesecond plurality of spread-spectrum signals. Accordingly, at the outputof the second plurality of matched filters is the second plurality ofdespread signals. Each matched filter of the second plurality of matchedfilters has an impulse response, h(t-T_(D1)), h (t-T_(o)-T_(D1)),h(t-2T_(o)-T_(D1)), etc., with a time delay T_(o) offset from the othermatched filters and with a time delay T_(D1) offset from the firstplurality of matched filters. A fourth matched filter 144 is coupledbetween the input and through the fourth weighting device 134 to thesecond adder 130. A fifth matched filter 145 is coupled between theinput, and through the fifth weighting device 135 to the second adder130. A sixth matched filter 146 is coupled between the input and throughthe sixth weighting device 136 to the second adder 130. As mentionedpreviously, the fourth weighting device 134, the fifth weighting device135, and the sixth weighting device 136 are optional. The fourthweighting device 134, the fifth weighting device 135, and the sixthweighting device 136, are coupled respectively to a source forgenerating the fourth weighting signal W₄, the fifth weighting signalW₅, and the sixth weighting signal W₆. Also, as with the correlatorembodiment, the first adder 120 and the second adder 130 are coupled tothe decision device 150. The decision device 150 may be embodied as aselector or a combiner.

The present invention may further include despreading spread-spectrum,signals located within a third group. Accordingly, the present inventionmay include third despreading means and third combining means. The thirdcombining means is coupled between the third despreading means and theselecting means.

As shown in FIG. 14, the third despreading means despreads the receivedsignal r(t) received as a third plurality or spread-spectrum signalswithin a third group. Accordingly, the third despreading means generatesa third plurality of despread signals. The third combining meanscombines the third plurality of despread signals as a thirdcombined-despread signal. The selecting means selects one of the firstcombined-despread signal, the second combined-despread signal or thethird combined-despread signal. The output of the selecting means is theoutput-despread signal.

As shown in FIG. 14, the third despreading means may include a thirdplurality of correlators for despreading the third plurality ofspread-spectrum signals. The third plurality of correlators isillustrated, by way of example, with seventh multiplier 117. eighthmultiplier 118, ninth multiplier 119, seventh filler 127, eighth filter128, ninth filter 129, and a source for generating the seventhchipping-sequence signal g(t-T_(D2)), the eighth chipping-sequencesignal g(t-T_(o)-T_(D2)), and the ninth chipping-sequence signalg(t-T_(o)-T_(D2)). The seventh multiplier 117 is coupled between theinput and the seventh filter 127. The eighth multiplier 118, is coupledbetween the input and the eighth filter 128. The ninth multiplier 119 iscoupled between the input and the ninth filter 129. The seventhmultiplier 117, the eighth multiplier 118, and the ninth multiplier 119,are coupled to the source for generating the seventh chipping-sequencesignal, the eighth chipping-sequence signal and the ninthchipping-sequence signal, respectively. Optionally, at the output of theseventh filter 127, eighth filter 128, and ninth filter 129, may beseventh weighting device 137, eighth weighting device 138, and ninthweighting device 139, respectively. Accordingly, the output of theseventh filter 127 is coupled through the seventh weighting device 137to the third adder 140. The output of the eighth filter 128 is coupledthrough the eighth weighting device 138 to the third adder 140. Theoutput of the ninth multiplier 129 is coupled through the ninthweighting device 139 to the third adder 140. The third adder is coupledto the decision device 150. At the output of the third plurality ofcorrelators is the third plurality of despread signals, respectively.

Preferably, each correlator of the third plurality of correlatorsdespreads with a chipping-sequence signal g(t-T_(D2)),g(t-T_(o)-T_(D2)), g(t-2T_(o)-T_(D2))having a time delay T_(o) differentfrom each time delay of each chipping-sequence signal used with othercorrelators of the third plurality of correlators. Also, each correlatorof the third plurality of correlators despreads with a chipping-sequencesignal having a time delay different from each time delay of eachchipping-sequence signal used, respectively, with each correlator of thesecond plurality of correlators. Also, each correlator of the thirdplurality of correlators despreads with a chipping-sequence signalhaving a time delay 2T_(D) different from each chipping-sequence signalused with each correlator of the first plurality of correlators.

Alternatively, the third despreading means may include, as shown in FIG.15, a third plurality of matched filters for despreading the thirdplurality of spread-spectrum signals. The third plurality of matchedfilters includes seventh matched filter 147, eighth matched filter 148,and ninth matched filter 149. The seventh matched filter is coupledbetween the input and through the seventh weighting device 137 to thethird adder 140. The eighth matched filter 148 is coupled between theinput and through the eighth weighting device 138 to the third adder140. The ninth matched filter 149 is coupled between the input andthrough the ninth weighting device 139 to the third adder 140. The thirdadder 140 is coupled to the decision device 150. At the output of thethird plurality of matched filters is the third plurality of despreadsignals.

The present invention may include fourth despreading means and fourthcombining means, with the fourth combining means coupled between thefourth despreading means and the selecting means. The fourth despreadingmeans would despread a fourth plurality of spread-spectrum signalswithin a fourth group. The output of the fourth despreading means wouldbe a fourth plurality of despread signals. The fourth combining meanswould combine the fourth plurality of despread signals as a fourthcombined-despread signal. The selecting means selects one of the firstcombined-despread signal, the second combined-despread signal, the thirdcombined-despread signal, or the fourth combined-despread signal, as theoutput-despread signal.

In a similar fashion, the fourth despreading means includes a fourthplurality of correlators, or a fourth plurality of matched filters, fordespreading the fourth plurality of spread-spectrum signals forgenerating the fourth plurality of despread signals. Each correlator ofthe fourth plurality of correlators would despread with achipping-sequence signal having a time delay different from each timedelay of each chipping-sequence signal used, respectively, with othercorrelators of the fourth plurality of correlators. Also, thechipping-sequence signal would be different from the chipping-sequencesignals used with each correlator of the third plurality of correlators,each chipping-sequence signal used with each correlator of the secondplurality of correlators, and each chipping-sequence signal used witheach correlator of the first plurality of correlators. Based on thedisclosure herein, a person skilled in the art would readily know how toextend the concept to a fifth group or spread-spectrum signals, or moregenerally, to a plurality of groups of spread-spectrum signals.

Each of the matched filters may be realized using surface-acoustic-wave(SAW) devices, digital matched filters, or embodied in an applicationspecific integrated circuit (ASIC) chip or a digital signal processor(DSP) chip. Techniques for designing matched filters using these devicesare well known in the art.

A multipath processor can single out individual paths from a group ofrays. The weight for each weighting device is figured out by sets ofcorrelators, and with a reference code it is possible to track thechipping-sequence signal in each ray.

Alternatively, a method using a multipath processor may be used fortracking a spread-spectrum signal within a plurality of groups. Themethod comprises the steps of despreading the received signal r(t)received as the first plurality of spread-spectrum signals within afirst group to generate a first plurality of despread signals. The firstplurality of despread signals are then combined as a firstcombined-despread signal. The method would include despreading thereceived signal r(t) received as a second plurality of spread-spectrumsignals within a second group to generate a second plurality of despreadsignals. The second plurality of despread signals would be combined as asecond combined-despread signal. The method includes selecting eitherthe first combined-despread signal or the second combined-despreadsignal, as an output-despread signal.

The step of despreading the first plurality of spread-spectrum signalsmay include the step of correlating or matched filtering the firstplurality of spread-spectrum signals, using a first plurality ofcorrelators or a first plurality of matched filters, respectively. Thestep of despreading the second plurality of spread-spectrum signals mayinclude the step of correlating or matched filtering the secondplurality of spread-spectrum signals using a second plurality ofcorrelators or a second plurality of matched filters, respectively.

The method may further include despreading a third plurality ofspread-spectrum signals within a third group to generate a thirdplurality of despread signals. The third plurality of despread signalswould be combined as a third combined-despread signal. The selectingstep would thereby include selecting one of the first combined-despreadsignal, the second combined-despread signal or the thirdcombined-despread signal, as the output-despread signal. Similarly, thestep of despreading the third plurality of spread-spectrum signals mayinclude the step of correlating or matched filtering the third pluralityof spread-spectrum signals using a third plurality of correlators or athird plurality of matched filters, respectively.

The step of despreading each of the first plurality of spread-spectrumsignals would include the step of despreading with a chipping-sequencesignal having a time delay different from each time delay of eachchipping-sequence signal used to despread other spread-spectrum signalsof the first plurality of spread-spectrum signals. Similarly, the stepof despreading each of the second plurality of spread-spectrum signalswould include the step of despreading with a chipping-sequence signalhaving a time delay different from each time delay of eachchipping-sequence signal used to despread other spread-spectrum signalsof the second plurality of spread-spectrum signals. Also, the step ofdespreading each of the second plurality of spread-spectrum signalswould include the step of despreading with a chipping-sequence signalhaving a time delay different from each time delay of eachchipping-sequence signal used to despread other spread-spectrum signalsof the first plurality of spread-spectrum signals.

In the event the method includes the step of despreading a thirdplurality of spread-spectrum signals, the method would include the stepsof despreading with a chipping-sequence signal having a time delaydifferent for each time delay of each chipping-sequence signal used todespread other spread-spectrum signals of the third plurality ofspread-spectrum signals. Also, the time delay would be different foreach chipping-sequence signal used to despread spread-spectrum signalsof the second plurality of spread-spectrum signals, and different fromeach time delay of each, chipping-sequence signal used to despreadspread-spectrum signals of the first plurality of spread-spectrumsignals.

The method may be extended to a fourth, fifth or plurality of groups ofspread-spectrum signals.

The present invention also includes a variable-bandwidth spread-spectrumdevice for use with a spread-spectrum transmitter. Thevariable-bandwidth spread-spectrum device generates a spread-spectrumsignal having a spread bandwidth. The term “spread bandwidth”, as usedherein, denotes the bandwidth of the transmitted spread-spectrum signal.The variable-band width spread-spectrum device uses a chipping-sequencesignal having a chipping rate which is less than the spread bandwidth.The term “chipping rate”, as used herein, denotes the bandwidth of thechipping-sequence signal.

The variable-bandwidth spread-spectrum device includes first generatingmeans, second generating means, spread-spectrum processing means, andfiltering means. The spread-spectrum processing means is coupled to thefirst generating means. The second generating means is coupled betweenthe spread-spectrum processing means and the filtering means.

The first generating means generates the chipping-sequence signal withthe chipping rate. The spread-spectrum processing means processes a datasignal with the chipping-sequence signal to generate a spread-datasignal. The second generating means generates an impulse signal, inresponse to each chip of the spread-data signal. The filtering meansfilters the spectrum of each, impulse signal with a bandpass equal tothe spread bandwidth.

As illustratively shown in FIG. 16, the first generating means may beembodied as a chipping-sequence generator 161, the second generatingmeans may be embodied as an impulse generator 165, the spread-spectrumprocessing means may be embodied as an EXCLUSIVE-OR gate product device164, or other device known to those skilled in the art for mixing a datasignal with a chipping-sequence signal, and the filtering means may beembodied as a filter 166.

The product device 164 is coupled to the chipping-sequence generator161. The impulse generator 166 is coupled between the product device 164and the filter 166.

The chipping-sequence generator 161 generates the chipping-sequencesignal with the chipping rate. The product device 164 processes the datasignal with the chipping-sequence signal, thereby generating aspread-data signal as shown in FIG. 17. The impulse generator 165generates an impulse signal, as shown in FIG. 18, in response to eachchip in the spread-data signal shown in FIG. 17. Each impulse signal ofFIG. 18 has an impulse bandwidth. The term “impulse bandwidth”, as usedherein, denotes the bandwidth of the impulse signal. While theoreticallyan impulse signal has infinite bandwidth, practically, the impulsesignal has a bandwidth which is greater than the spread bandwidth.

The filter 166 has a bandwidth adjusted to the spread bandwidth. Thus,the filter 166 filters a spectrum of each impulse signal of thespread-data signal with the spread bandwidth. The filter 166 does thisfor each impulse signal.

The filter 166 preferably includes a variable-bandwidth filter. Thevariable-bandwidth filter may be used for varying or adjusting thespread bandwidth of the spectrum for each impulse signal. Accordingly, aspread-spectrum signal may be designed having the bandwidth of choice,based en the bandwidth of the variable-bandwidth filter. The bandwidthmay be variable, or adjustable, as would be required for particularsystem requirements. As used in this patent, a variable bandwidth is onethat is able to vary according to time conditions, background signals orinterference, or other requirements in a particular system. Anadjustable bandwidth would be similar to a variable bandwidth, but isused to refer to a bandwidth which may be adjusted to remain at a chosensetting.

The first generating means, as shown in FIG. 19, may include afrequency-domain-chipping-sequence generator 161 and aninverse-Fourier-transform device 162. Thefrequency-domain-chipping-sequence generator 161 may be used to generatea frequency-domain representation of a chipping-sequence signal. Theinverse-Fourier-transform device 162 transforms the frequency-domainrepresentation of the chipping-sequence signal to the chipping-sequencesignal.

The first generating means may further include a memory 163 for storingthe chipping-sequence signal.

The present invention also includes a variable-bandwidth spread-spectrummethod for use with a transmitter. The method includes the steps ofgenerating the chipping-sequence signal with the chipping rate, andspread-spectrum processing a data signal with, the chipping-sequencesignal to generate a spread-data signal. Each chip in thespread-spectrum signal is used to generate an impulse signal. Eachimpulse signal is filtered with the spread bandwidth to generate thedesired bandwidth signal.

Thus, the variable-bandwidth-spread-spectrum device uses a lower chiprate, but provides a wider bandwidth signal. The power spectral densityat the output of the filter 166 of the filtered-spread-data signal s(t)is proportional to the frequency response H(f) of the filter.

Equation 3

PSD _(s(t)) =k|H(f)|²   i.

Thus, the filter 166 controls the shape of the spectrum of thefiltered-spread-data signal.

The processing gain (PG) is bandwidth W of the filtered-spread-datasignal divided by chip rate Rb of the filtered-spread-data signal.

Equation 4

PG=W/R _(b)   i.

The capacity N of the filter-spread-data signal is

$\begin{matrix}{{Equation}\mspace{14mu} 5} & \; \\{N \leq {\frac{PG}{E_{b}/N_{o}} + 1.}} & {ii}\end{matrix}$

The capacity does not depend on chip rate, but instead on bandwidth. Onecan achieve an upper bound on the capacity if the chip rate is greaterthan the bandwidth. But, if the chip rate were lower, then one can savepower consumption, i.e. use a lower clock rate of CMOS, which determinespower consumption.

The present invention assumes that a plurality of mobile stationsoperate in a cellular-communications network using spread-spectrummodulation. The cellular communications network has a plurality ofgeographical regions, with a multiplicity of cells within eachgeographical region. The size of the cells in a first geographicalregion may differ from the size of the cells in a second geographicalregion. In a first geographical region, such as an urban environment,the cellular architecture may have a large number of cells, each ofsmall area, which place the corresponding base station, close to eachother. In a second geographical region, such as a rural environment, thecellular architecture may have a smaller number of cells, each of largerarea. Further, the size of the cells may vary even within a specifiedgeographic region.

A mobile station, while in the urban environment of the firstgeographical region, may be required to transmit at a lower power levelthan while in the rural environment of the second geographical region.This requirement might be due to a decreased range of the mobile stationfrom the base station. Within a particular cell, a mobile station nearthe base station of the cell may be required to transmit with a powerlevel less than that required, when the mobile station is near an outerperimeter of the cell. This adjustment in power level is necessary toensure a constant power level is received at the base station from eachmobile station.

Adaptive power control works by measuring the received signal to noiseratio (SNR) for each user and causing the user transmitted power to varyin a manner to cause all users' SNR's to be equal to a common valuewhich will be adequate for reliable communication if the total number ofusers and interference is less than system capacity. While this assumesthat all users are obtaining the same service, e.g., 32 kbs voice data,it is a feature of the system described that different service optionsare supported for requesting users. This is done by adjusting thesetpoint for each user independently.

There are two issues that arise when addressing the base operation of anadaptive power control system. The first is the common value obtainedfor SNR versus the load and its cost to the transmitters in terms oftransmitted power, and the second is the stability of the system.Stability means that a perturbation of the system from its quiescentstate causes a reaction of the system to restore the quiescentcondition. It is highly desirable that there exist only one quiescentpoint because otherwise “chatter” or oscillation may occur. Stabilitymust be addressed with any control system but, in the present case, thesituation is somewhat complicated by the fact that the users affect oneanother, and thereby cause the control variables, the transmitted powerand resulting SNB's, to be dynamically coupled. The coupling is apparentwhen one realizes that all signals are processed by a common AGCfunction which does not discriminate individual user signals from eachother or from other sources.

The power control scheme of the present invention is a closed loopscheme. The system measures the correlator output power for each userand compares the measured value with a target value or setpoint. Thismeasured power includes both the desired signal component and unwantedpower or noise.

The AGC maintains the total power into each correlator at a presetlevel. This level does not vary as a function of APC action; that is,this role of the AGC is independent of APC. Furthermore, an increase inreceived power from any user or subset of users will be “attacked” bythe AGC. This is possible because the AGC time constant is smaller thanthe APC time constant, i.e., the AGC is faster than the APC. Since thetotal power available out of the AGC is fixed, an increase in theportion due to one user comes at the expense of all other users. Whilethis may work against the apparent stability of the system, the AGCsensor, which measures the AGC control signal and thereby measures thetotal received power, causes the system to seek a quiescent statecorresponding to the minimum received power per user. It is desired thatthe transmitted power be minimized because this will minimize intercellinterference and conserve battery power. Excess transmitter power willbe dissipated within the AGC as long as all users transmit excessivepower.

The implementation shown, in the figures is to be consideredrepresentative. In particular, the method of controlling the remotetransmitter power via attenuators and variable gain amplifiers isperhaps redundant. Either or both of these means may be employed,depending upon the (dynamic) range of control required. Also, controlmay be caused at either IF or RF frequencies.

For discussion purposes, a mobile station within a particular celltransmits a first spread-spectrum signal, and the base station transmitsa second spread-spectrum signal. In the exemplary arrangement shown inFIG. 20, a block diagram of a base station as part of a system foradaptive-power control of a spread-spectrum transmitter is provided.

FIG. 20 illustrates the base station adaptive power control system, withautomatic gain control (AGC) means, power means, comparator means,transmitter means, and an antenna. The AGC means is shown as anautomatic-gain-control (AGC) amplifier 228, correlator means is shown asdespreader 231, and power moans is shown as power measurement device233. The comparator means is shown as comparator 239, the transmittermeans is shown as power amplifier 237 coupled to the antenna 226. Alsoillustrated is a delta modulator 235 coupled between comparator 239 andpower amplifier 237.

The AGC amplifier 228 is coupled to the despreader 231. The powermeasurement device 233 is coupled to the despreader 231. The comparator239 is coupled to the output of the power measurement device 233 and tothe AGC amplifier 228. The multiplexer 234 is coupled between thecomparator 239 and the power amplifier 237. The delta modulator 235 iscoupled between the power amplifier 237 and the multiplexer 234. Thepower amplifier 237 is coupled to the antenna 226.

A threshold level is used by the comparator 239 as a comparison for thereceived-power level measured by the power measurement device 233.

For each received signal, the AGC amplifier 228 generates an AGC-outputsignal and an AGC-control signal. The AGC-output signal is despread toobtain the signal of a first user using despreader 231. Thedespread-AGC-output signal from the despreader 231 is combined with theAGC-control signal from the AGC amplifier 228, by the combiner 241. TheAGC-control signal from the AGC amplifier 228 may be offset by offsetlevel S₁ using combiner 242, and weighted by weighting device 243. Theweighting device 243 may be an amplifier or attenuator.

The received-power level from power device 233 may be offset by offsetlevel S₂ using combiner 244, and weighted by weighting device 245. Theweighting device 245 may be an amplifier or attenuator. The combiner 241combines the AGC-control signal with the received-level signal, forgenerating adjusted-received-power level. The comparator 239 generates acomparison signal by comparing the adjusted-received-power level to thethreshold level. The comparison signal may be an analog or digital datasignal. The comparison signal indicates whether the mobile station is toincrease or decrease its power level. If the adjusted-received-powerlevel exceeds the threshold, for example, then the comparison signalsends a message to the mobile station to decrease its transmitter power.If the adjusted-received-power level were below the threshold, then thecomparison signal sends a message to the mobile station to increase itstransmitter power. The comparison signal is converted to a power-commandsignal by the delta modulator 235.

The power-command signal may be transmitted with or separate from thesecond spread-spectrum signal. For example, a spread-spectrum signalusing a first chip sequence may be considered a first spread-spectrumchannel, end a spread-spectrum signal using a second chip sequence maybe considered a second spread-spectrum channel. The power-command signalmay be transmuted in the same spread-spectrum channel, i.e., the firstspread-spectrum channel, as the second spread-spectrum signal, in whichcase the power-command signal is transmitted at a different timeinterval from when the second spread-spectrum signal is transmitted.This format allows the mobile station to acquire synchronization withthe first sequence, using the second spread-spectrum signal. Thepower-command signal may also be transmitted in a second spread-spectrumchannel which is different from the second spread-spectrum signal. Inthis case, the second spread-spectrum signal having the power-commandsignal would be acquired by the second chipping-sequence generator andsecond product device. The power-command signal may be time divisionmultiplexed or frequency division multiplexed with the secondspread-spectrum signal.

The base-correlator means is depicted in FIG. 20 as first despreader231. The system, by way of this example, may have the base-correlatormeans embodied as a product device, a chip-sequence generator, and abandpass filter. Alternatively, the base-correlator means may berealized as a matched filter such as a surface-acoustic-wave device, oras a digital matched filter embodied in a digital signal processor. Ingeneral, the base-correlator means uses or is matched to the chipsequence of the spread-spectrum signal being received. Correlators andmatched filters for despreading a spread-spectrum signal are well knownin the art.

Typically, the AGC circuit 228 is coupled to a low noise amplifier 227,through an isolator 225 to the antenna 226. In FIG. 20 a plurality ofdespreaders, despreader 229 through despreader 231, are shown fordespreading a plurality of spread-spectrum channels which may bereceived from a plurality of mobile stations. Similarly, the output ofeach despreader 229 through despreader 231 is coupled to a plurality ofdemodulators, illustrated as demodulator 230 through demodulator 232,respectively, for demodulating data from the despread AGC-output signal.Accordingly, a plurality of data outputs are available at the basestation.

For a particular spread-spectrum channel, the first despreader 231 isshown coupled to power device 233 and multiplexer 234. The power device233 typically is a power-measurement circuit which processes thedespread AGC-output signal as a received-power level. The power device233 might include an analog-to-digital converter circuit for outputtinga digital received-power level. The comparator means, embodied ascomparator circuit 239, compares the processed received-power level to athreshold. The multiplexer 234 is coupled to the output of the powerdevice 233 through the comparator circuit 239. The multiplexer 234 mayinsert appropriate framing bits, as required.

The transmitter means may be embodied as a quadrature phase shift keying(QPSK) modulator or a delta modulator 235 coupled to a power amplifier237. In FIG. 20, the input to the delta modulator 235 typically wouldhave the comparison signal from the comparator 239 multiplexed with datafrom the k^(th) channel. The delta modulator 235 converts the comparisonsignal to a power-command signal. A plurality of spread spectrumchannels would have their data and appropriate power-command signalscombined by combiner 236 and amplified by power amplifier 237. Theoutput of the power amplifier 237 is coupled through the isolator 125 tothe antenna 226.

The power command signal is transmitted periodically. The period T mightbe chosen to be 250 microseconds in order to ensure a low root meansquare error as well as a low peak error between the instantaneousreceived signal and the constant desired signal.

A mobile station is illustratively shown in FIG. 21. Themobile-despreading means is illustrated as despreader 334 andvariable-gain means is illustrated as a variable-gain device 341. Thevariable-gain device 341 is coupled between the transmitter 342 andthrough isolator 336 to antenna 335. The despreader 334 is coupled tothe isolator 336 and to demultiplexer 339. The output of the despreader334 is also coupled to a demodulator 340. The despreader 334 may beembodied as an appropriate correlator, or matched filter, fordespreading the k^(th) channel. Additional circuitry may be used, suchas radio frequency (RF) amplifiers and filters, or intermediatefrequency (IF) amplifiers and filters, as is well known in the art.

A received second spread-spectrum signal at antenna 335 passes throughisolator 336 to despreader 334. The despreader 334 is matched to thechip sequence of the desired spread-spectrum channel. The output of thedespreader 334 passes through the demodulator 340 for demodulating thedata from the desired spread-spectrum channel. Additionally, thedemultiplexer 339 demultiplexes the power-command signal from thedespread signal outputted from despreader 334. The power-command signaldrives the variable-gain device 341.

A decision device 345 and accumulator 346 may be coupled between thedemultiplexer 339 and the variable gain device 341. Astep-size-algorithm device 344 is coupled to the output of the decisiondevice 345 and to the accumulator 346.

The step-size-algorithm device 344 stores an algorithm for adjusting thepower level of the variable gain device 341. A nonlinear algorithm thatmight be used is shown in FIG. 22. FIG. 23 compares a nonlinearalgorithm with a linear step size algorithm.

The power-command signal from the demultiplexer 339 causes the decisiondevice 345 to increase or decrease the power level of the variable gaindevice 341, based on the threshold of the step size algorithm shown inFIG. 23. The accumulator tracks previous power levels as a means forassessing the necessary adjustments in the step size of the power levelpursuant to the algorithm as shown in FIG. 23.

The variable-gain device 341 may be embodied as a variable-gainamplifier, a variable-gain attenuator, or any device which performs thesame function as the variable-gain, device 341 as described herein. Thevariable-gain device 341 increases or decreases the power level of theremote station transmitter, based on the power-command signal,

As illustratively shown in FIG. 20, a block diagram of a powermeasurement circuit includes interference rejection for use with thebase station. As shown in FIG. 20, the AGC amplifier 228 is connected tothe despreader 231, and the output of the despreader 231 is connected tothe power measurement circuit 233. Additionally, the AGC amplifier 228is connected to the combiner 236 through the comparator 239.

A received signal includes a first spread-spectrum signal with powerP_(C) and the other input signals which are considered to be interferingsignals with power P_(J) at the input to the AGC amplifier 228 of FIG.20. The interfering signal may come from one or more nondesirablesignals, noise, multipath signals, and any other source which wouldserve as an interfering signal to the first spread-spectrum signal. Thereceived signal is normalized by the AGC amplifier 228. Thus, by way ofexample, the AGC amplifier 228 can have the power output, P_(C)+P_(J)=1.The normalized received signal is despread by the despreader 231 toreceive a particular mobile user's signal. The chipping-sequencegenerator of despreader 231 generates a chip-sequence signal using thesame chip sequence as used by the first spread-spectrum signal.Alternatively, the matched filter, if used, of despreader 231 may havean impulse response matched to the same chip sequence as used by thefirst spread-spectrum signal. The output of the despreader 231 is thenormalised power or the first spread-spectrum signal plus the normalizedpower or the interfering signal divided by the processing gain, PG, ofthe spread-spectrum system. The power measurement circuit 233 generatesa received-power level of the first spread-spectrum signal. Thecomparator 239 processes the despread-received signal with theAGC-control signal and outputs the power-control signal of the firstspread-spectrum signal. The power level of the interfering signal isreduced by the processing gain, PG.

The comparator 239 processes the AGC-control signal with the despread,normalized received signal, by multiplying the two signals together, orby logarithmically processing the AGC-control signal with thedespread-received signal. In the latter case, the logarithm is taken ofthe power of the received signal, P_(C)+P_(J), and the logarithm istaken of the despread, normalized received signal. The two logarithmsare added together to produce the received-power level.

For the present invention to work effectively, the despread signal mustbe kept nearly constant, independent of variations in the other signalsor of obstructions. A preferred implementation to accomplish this end isshown in the circuitry of FIG. 20. FIG. 20 depicts a omens fordetermining at the base station the power of the first spread-spectrumsignal when the received signal includes multiple signals and noise. Ifthe circuitry of FIG. 20 were not used, then it is possible that theinterfering signal, which may include noise, multipath signals, andother undesirable signals, may raise the power level measured at theinput to the receiver of the base station, thereby suppressing the firstspread spectrum signal. The undesirable power Level measured may causethe remote station to transmit more power than required, increasing theamount of power received at the base station.

As noted earlier, the APC system is a closed loop system. The AFC loopoperates by generating commands to increase or decrease the transmitterpower at the update rate. This is actually quantization process that isdone to limit the amount of information that must be fed back to theremote transmitter. The amount of increase or decrease may be fixed inadvance or it may adapt in response to the characteristics of thechannel as measured locally in the remote terminal, the terminal beingcontrolled. In particular, the remote terminal may examine the sequenceof commands received by it. A long sequence of increase commands, forexample, implies that the step size may be increased. A typical schemeincreases the step size by a fixed amount or a fixed percentage whenevertwo successive bits are the same. For example, the step size may beincreased by 50% if two bits in a row are the same and decreased by 50%if they differ. This is a fairly gross change in the step size, and isintended to be adaptive to local, or immediate in time, variations inthe required transmitted power. This process results in a largevariation of the step size with time.

An adaptive step size algorithm may also be considered in a differentcontest. Specifically, the step size may be considered to be nearlyconstant or not responding to localized variations in demandedtransmitted power, but the value may be automatically adjusted based onthe global characteristics of the channel induced control action. Thus,in a nearly static environment one should use a small constant step sizewhile in a mobile environment the step size should be larger.

Adjustment of the power level of the remote station transmitter may beeffected either linearly or nonlinearly. The following algorithm willcause the step size to settle at a nearly optimum constant value. Thereceiver examines successive APC bits and increases the step size by thefactor (1+x) if they agree and decreases the step size by the factor(1+x) if they disagree. Here the parameter x is small (x=0.01, orexample). While this procedure will not allow local adaptation (becausex is small), it will result in an adaptation to global conditions.Specifically, if the transmitted APC bit stream exhibits a tendencytoward successive bits in agreement (i.e., runs of 1's or 0'sare-evident) it implies that the system is not following the changes inchannel conditions (i.e., the system is slow rate limited) and the stepsize should be increased. On the other hand, if successive bits tend tobe opposite, the system is “hunting” for a value between two values thatare excessively far apart. The statistics one expects to observe addsoptimal are intermediate to these extremes. That is, the ADC bit streamshould appear equally likely to contain the patterns (0,0), (0,1),(1,0), and (1,1) in any pair of successive bits. The above algorithmdrives the system behavior toward this.

The above algorithm (global adaptation) works particularly well when thesystem employs a high update rate relative to the dynamics of thechannel.

As illustrated in FIG. 23, to increase the power level using linearadjustment, for example, the transmitter power is increased in regularincrements of one volt, or other unit as instructed by the base station,until the power level received at the base station is sufficientlystrong. Linear adjustment may be time consuming if the power adjustmentnecessary were substantial.

As shown in FIG. 22, to increase the power using nonlinear adjustment,the transmitter voltage may be increased, by way of example,geometrically until the transmitted power is in excess of the desiredlevel. Transmitter power may be then reduced geometrically untiltransmitted power is below the desired level. A preferred approach is toincrease the step size voltage by a factor of 1.5 and to decrease thestep size by a factor of 0.5. Other nonlinear algorithms may be used. Asshown in FIG. 23, this process is repeated, with diminishing margins oferror in both excess and insufficiency of desired power, until thedesired signal level has been obtained. Nonlinear adjustment provides asignificantly faster rise and fall time than does linear adjustment, andmay be preferable if power must be adjusted significantly.

The system determines the error state (APC bit) every T sections, 1/Tbeing the update rate of the control. The update rate may vary from 100Hz, which is low, to 100 kHz, which is quite high. The opportunity tomeasure the error state of the system arises with each reception of anew symbol. Thus, the update rate may be equal to the symbol rate. Ifsuch an update rate is not supported, it is beneficial to make use ofthe available error measurements by combining them (or averaging them)between updates. This minimizes the chance of causing a power adjustmentin the wrong direction which can occur because of noise in the errorsignals themselves.

The choice of update rate depends on factors other than APC operation,namely, the amount of capacity and method of allocating capacity to thetransport of the APC bits over the channel. In general, a faster updatewill produce superior performance, even if the increased update rate isobtained by permitting the APC bits to be received in erroroccasionally. Elaborating, a 1 kHz update rate with no channel inducederrors will perform less effectively than a 100 kHz update rate at a 25%rate of errors. This is because of the self correcting behavior of thecontrol loop. A faster update rate eliminates the latency of controlwhich is a key performance limiting phenomenon.

A spread spectrum base station receives all incoming signalssimultaneously. Thus, is a signal were received at a higher power levelthan the others, then that signal's receiver has a highersignal-to-noise ratio and therefore a lower bit error rate. The basestation ensures that each mobile station transmits at the correct powerlevel by telling the remote, every 500 microseconds, whether to increaseor to decrease the mobile station's power.

FIG. 24 shows a typical fading signal which is received at the basestation along with ten other independently fading signals and thermalnoise having the same power as one of the signals. Note that the fadeduration is about 5 milliseconds which corresponds to vehicular speedexceeding 60 miles per hour. FIGS. 25-36 illustrate the results obtainedwhen using a particular adaptive power control algorithm. In this case,whenever the received signal changes power, the base station informs theremote and the remote varies its power by ±1 dB. FIG. 25 show theadaptive power control signal at the remote station. FIG. 26 shows thereceived power at the base station. Note that the adaptive power controltrack the deep fades and as a result 9 dB fades resulted. This reducedpower level resulted in a bit error of 1.4×10⁻².

For the same fade of FIG. 24, assume a different adaptive power controlalgorithm is employed as shown in FIGS. 27-28. In this case the controlvoltage results in the remote unit changing its power by a factor of 1.5in the same direction, or by a factor of 0.5 in the opposite direction.In this particular implementation the minimum step size was 0.25 dB andthe maximum step size was 4 dB. Note that the error is usually limitedto +0.2 dB with occasional decreases in power by 5 dB to 6 dB resultingin a BER≈8×10⁴, a significant improvement compared to the previousalgorithm. The use of interleaving and forward error correcting codesusually can correct any errors resulting from the rarely observed powerdips.

In operation, a mobile station in a cell may transmit the firstspread-spectrum signal on a continuous basis or on a repetitive periodicbasis. The base station within the cell receives the firstspread-spectrum signal. The received first spread-spectrum signal isacquired and despread with the chip-sequence signal from chip-sequencegenerator and product device. The despread first spread-spectrum signalis filtered through bandpass filter. The base station detects thedespread first spread-spectrum signal using envelope detector, andmeasures or determines the received-power level of the firstspread-spectrum signal. The base station generates the power-commandsignal from the received-power level.

The present invention also includes a method for automatic-power controlof a spread-spectrum transmitter for a mobile station operating in acellular-communications network using spread-spectrum modulation, withthe mobile station transmitting a first spread-spectrum signal. In use,the method includes the step of receiving a received signal, generatingan AGC-output signal, despreading the AGC-output signal, processing thedespread AGC-output signal to generate a received-power level,generating a power-command signal, transmitting the power-command signalas a second spread-spectrum signal, despreading the power-command signalfrom the second spread-spectrum signal as a power-adjust signal, andadjusting a power level of the first spread-spectrum signal.

The received signal includes the first spread-spectrum signal and aninterfering signal and is received at the base station. The AGC-outputsignal is generated at the base station and despread as a despreadAGC-output signal. The despread AGC-output signal is processed at thebase station to generate a received-power level.

The received-power level is compared to a threshold, with the comparisonused to generate a power-command signal. If the received-power levelwere greater than the threshold, the power-command signal would commandthe mobile station to reduce transmitter power. If the received-powerlevel were less than the threshold, the power-command signal wouldcommand the mobile station to increase transmitter power.

The power-command signal is transmitted from the base station to themobile station as a second spread-spectrum signal. Responsive toreceiving the second spread-spectrum signal, the mobile stationdespreads the power-command signal as a power-adjust signal. Dependingon whether the power-command signal commanded the mobile station toincrease or decrease transmitter power, the mobile station, responsiveto the power adjust signal, increases or decreases the transmitter-powerlevel of the first spread-spectrum signal respectively.

The method may additionally include generating from a received signal anAGC-output signal, and despreading the AGC-output signal. The receivedsignal includes the first spread-spectrum signal and an interferingsignal. The received signal is processed with the despread AGC-outputsignal to generate a received-power level. The method then generates acomparison signal by comparing the received-power level to the thresholdlevel. While transmitting a second spread-spectrum signal, the methodadjusts a transmitter-power level of the first spread-spectrum signalfrom the transmitter using the power-adjust signal.

It will be apparent to those skilled in the art that variousmodifications can be made to the spread-spectrum system and method ofthe instant invention without departing from the scope or spirit of theinvention, and it is intended that the present invention covermodifications and variations of the spread-spectrum system and methodprovided they come within the scope of the appended claims and theirequivalents.

1-12. (canceled)
 13. A mobile station comprising: at least one downlinkantenna configured to receive a plurality of transmit power controlcommands; and algorithm device circuitry configured to implement atleast two power control algorithms, including: a first power controlalgorithm to adjust a transmission power level by increasing ordecreasing the transmission power level by a same, equal amount for eachreceived transmit power control command to increase or decrease thetransmission power level, respectively; and a second power controlalgorithm to adjust the transmission power level based on a thresholdstep size of the second power control algorithm.
 14. The mobile stationof claim 13, further comprising a variable gain device configured toadjust the transmission power level in response to the algorithm devicecircuitry.
 15. The mobile station of claim 13, further compromising ade-spreader for de-spreading a received spread-spectrum channel, thereceived spread-spectrum channel carrying the plurality of transmitpower control commands, the plurality of transmit power control commandsto be multiplexed in the received spread-spectrum channel; and ademultiplexer configured to demultiplex the plurality of transmit powercontrol commands from the despread received spread-spectrum channel. 16.The mobile station of claim 13, wherein the threshold step size of thesecond power control algorithm is less than a minimum power control steplevel of the at least one downlink antenna.
 17. A method for adjusting atransmission power level by a mobile station comprising: receiving aplurality of transmit power control commands via at least one downlinkantenna; and implementing at least two power control algorithms,including: a first power control algorithm to adjust the transmissionpower level by increasing or decreasing the transmission power level bya same, equal amount for a received transmit power control command toincrease or decrease the transmission power level, respectively; and asecond power control algorithm to adjust the transmission power levelbased on a threshold step size of the second power control algorithm.18. The method of claim 17, further comprising adjusting thetransmission power level in response to the plurality of transmissionpower level control commands.
 19. The method of claim 17 furthercomprising: de-spreading a received spread-spectrum channel, thereceived spread-spectrum channel carrying the plurality of transmitpower control commands, the plurality of transmit power control commandsto be multiplexed in the received spread-spectrum channel; anddemultiplexing the plurality of transmit power control commands from thede-spread received spread-spectrum channel.
 20. The method of claim 17,wherein the threshold step size of the second power control algorithm isless than a minimum power control step level of the at least onedownlink antenna.
 21. A system comprising: a base station to transmit aplurality of transmit power control commands; and a mobile stationcomprising: at least one downlink antenna configured to receive theplurality of transmit power control commands from the base station; andalgorithm device circuitry configured to implement at least two powercontrol algorithms, including: a first power control algorithm to adjusta transmission power level by increasing or decreasing the transmissionpower level by a same, equal amount for each received transmit powercontrol command to increase or decrease the transmission power level,respectively; and a second power control algorithm to adjust thetransmission power level based on a threshold step size of the secondpower control algorithm.
 22. The system of claim 21, wherein the mobilestation further comprises a variable gain device configured to adjustthe transmission power level in response to the algorithm devicecircuitry.
 23. The system of claim 21, wherein the mobile stationfurther comprises a de-spreader for de-spreading a receivedspread-spectrum channel, the received spread-spectrum channel carryingthe plurality of transmit power control commands, the plurality oftransmit power control commands to be multiplexed in the receivedspread-spectrum channel; and a demultiplexer configured to demultiplexthe plurality of transmit power control commands from the despreadreceived spread-spectrum channel.
 24. The system of claim 21, whereinthe threshold step size of the second power control algorithm of thealgorithm device circuitry of the mobile station is less than a minimumpower control step level of the at least one downlink antenna of themobile station.